QAM spread spectrum demodulation system

ABSTRACT

A demodulation arrangement is disclosed which comprises a mixing circuit for deriving in-phase (I) and quadrature (Q) signals in a digital form from a received signal. The I and Q signals can then be input to a look-up table configured to output at least a data output value corresponding to a symbol constellation point and a corresponding phase output value, both selectable using. A phase processor is arranged to receive the phase output value and to calculate a phase angle value of the received signal. The phase angle value is returned as a further input to the look-up table to supplement the I and Q signals for selection of at least the data output value.

FIELD OF THE INVENTION

The present invention relates generally to radio frequency (RF)communication systems and, in particular, discloses a quadratureamplitude modulation QAM spread spectrum demodulation system suitablefor general application, but also finding particular application inwireless local area networks.

BACKGROUND ART

Associated with increases in office automation based around computernetworks over the past ten years, there has developed an increasingdesire to reduce, and ultimately to eliminate, data cable connectionsbetween devices such as main frame computers, terminals, personalcomputers, printers and the like. This has resulted in the developmentof so-called "wireless local area networks" (wireless LAN's), whichutilise bi-directional radio frequency (RF) communications betweendevices arranged within such a system. Such systems present essentiallythree basic problems that have hampered their development. Firstly, inview of the number of individual number of equipments generally used insuch a system, the hardware cost of any RF transceiver to be associatedwith each item of equipment should be minimized to ensure costeffectiveness. Further, the desired data rate, which is currentlyapproximately 2 megabits per second (Mbps), and which is anticipated toincrease to approximately 10 Mbps by the year 2000 generallynecessitates microwave transmission frequencies, which are typicallyused for point-to-point communications over substantially largerdistances. The third requirement is that of transmitter power. On thislast point, the transmitted power level must be sufficiently low to fallwithin acceptable limits of environmental radiation hazards associatedwith office work places, and yet have sufficient power to achieveadequate propagation.

QAM spread spectrum systems when implemented at a high level, forexample 64 QAM, have the advantages of being able to provide a highspectral efficiency which allow data rates in excess of 10 Mbps to beachieved whilst meeting the processing gain requirements established byUnited States Regulation FCC15.247, and a similar Japanese regulation,in a bandwidth of less than 26 MHz. Also, such systems provide animproved peak/average power ratio relative to other known transmissionsystems.

However, such systems have disadvantages. In particular, the traditionalimplementation of 64 QAM demodulation with a symbol rate of around 2Mbps is complex, and as a result, not generally suited to the size andpower requirement necessary for wireless LANs. Further, the low distancebetween symbols in a 64 QAM constellation causes susceptibility tomulti-path effects, otherwise known as frequency selective fading.Additionally, 64 QAM is highly susceptible to amplitude modulation tophase modulation (AM/PM) conversion that occurs within the poweramplifier of the RF transmitter. Traditional techniques to overcome thisproblem include designing expensive, highly linear power amplifiers, oralternatively operating power amplifiers approximately 10 dB below therated output. Such techniques generally do not meet thecost-requirements for wireless LAN applications.

U.S. Pat. No. 5,005,186 (Aono et al) discloses a quadrature amplitudemodulation (QAM) spread spectrum demodulator which permits high datarate and relatively low cost implementation through the use of aread-only-memory (ROM) which operates for constellation decoding and toprovide error correction signals. However, such an arrangement isunderstood to be configured for microwave point-to-point applicationsand is generally unsuitable to wireless LAN environments.

Accordingly, it would be advantageous to provide a radio frequencytransmission system which at least ameliorates the deficienciesmentioned above and thereby permits implementation in wireless LANapplications.

SUMMARY OF THE INVENTION

In accordance with one aspect of the present invention there isdisclosed a demodulation arrangement comprising:

a mixing circuit for deriving in-phase (I) and quadrature (Q) signals ina digital form from a received signal,

a look-up table comprising a plurality of data output values and phaseoutput values and configured to output at least a data output valuecorresponding to a symbol constellation point and a corresponding phaseoutput value, both selectable using the I and Q signals, and

a phase processor arranged to receive the phase output value and tocalculate a phase angle value of the received signal, wherein

the phase angle value is further input to the look-up table tosupplement the I and Q signals for selection of at least the data outputvalue.

In accordance with another aspect of the present invention there isdisclosed a demodulation apparatus comprising a ROM in which data valuesand phase values are stored for selection by I and Q input data, a phaseprocessor for determining, from a plurality of the phase values, acurrent phase angle of the I and Q input data, and a feedback path bywhich the current phase angle is input directly to the ROM for modifyingthe I and Q values prior to the selection.

In accordance with another aspect of the present invention there isdisclosed a QAM demodulation apparatus comprising a ROM configured fordecoding symbol constellation points from a received signal,characterised in that a feedback path is formed from and returningdirectly to the ROM, the feedback path implementing a digital carrierrecovery loop incorporating a complex multiplication function.

In accordance with another aspect of the present invention there isdisclosed a method of demodulating a modulated signal, the methodcomprising the steps of:

(a) deriving in-phase (I) and quadrature (Q) signals from the signal;

(b) obtaining a decoded data output value and a phase output valuecorresponding to the I and Q signals;

(c) determining a phase angle value of the signal from the phase outputvalue; and

(d) using the phase angle value in concert with the I and Q signals inthe obtaining of at least the decoded data output value.

BRIEF DESCRIPTION OF THE DRAWINGS

A number of aspects of the background art, and a number of preferredembodiments of the present invention will now be described withreference to the drawings in which:

FIG. 1 is a schematic block diagram representation of an RF transmissionsystem of conventional structure, and also which permits implementationusing apparatus configured in accordance with the preferred embodiment;

FIG. 2 is a schematic block diagram representation of a local wirelessLAN in which the preferred embodiment can be used;

FIGS. 3A to 3C illustrate respectively QPSK (4 QAM), 16 QAM and 64 QAMsignal constellations;

FIG. 3D illustrates the effects of AM/PM conversion on one quadrant ofthe constellation of FIG. 3C;

FIG. 3E is a phase plot of AM/PM conversion for a typical RF poweramplifier;

FIG. 4 illustrates a prior art QAM demodulator which incorporatescarrier signal recovery;

FIG. 5 is a schematic block diagram representation of an RF receiver ofone embodiment;

FIG. 6 is a schematic block diagram representation of a QAM demodulatorwhich can be used in the receiver of FIG. 5 and which incorporates a QAMdigital signal processor (DSP);

FIG. 7 is a schematic block diagram representation of the preferred QAMprocessing function which can be implemented in the processor of FIG. 6;

FIG. 8 is a schematic block diagram representation of a preferredembodiment of the function of FIG. 7;

FIG. 9 is a schematic block diagram representation of an alternativeembodiment of the function illustrated in FIG. 7;

FIG. 10 illustrates a preferred transmission packet configuration foruse with the preferred embodiment;

FIG. 11 illustrates a QAM demodulator of another embodiment whichutilises pre-processing; and

FIG. 12 is a schematic block diagram representation of an alternativeembodiment which uses post processing.

DETAILED DESCRIPTION

FIG. 1 shows a communication system 1 configured for point-to-pointcommunication between a transmitter 2 arranged at one location and areceiver 8 located at another. The transmitter 2 incorporates amodulator 3 which receives input data 6 and which outputs to an RF poweramplifier 4 that is coupled to an antenna 5. The antenna 5 emits atransmitted signal 7 which is detected by an antenna 9 coupled to aradio frequency receiver 10 incorporated within a receiver 8. The RFreceiver 10 outputs to a demodulator 11 which reproduces output data 12from the transmitted signal.

The system 1 can be configured with any form of modulation systemhowever, when used for microwave point to point communications, QAM isoften used in view of it providing a high data rate. Furthermore, insuch applications the relative cost of equipment is small compared tothe distances travelled by communications signals and the volume oftraffic that can be carried.

FIG. 2 illustrates a wireless LAN 15 which incorporates transceivermodules 20 each configured with a transmitter 2 and receiver 8 of thearrangement of FIG. 1. In this case, the modules 20 incorporate acontroller 21 which receives bi-directional data 22 and provides thedata input 6 to the transmitter 2, and receives the data output 12 fromthe receiver 8. The transmitter 2 and receiver 8 each couple to a RFtransceiver terminal 24 to which an antenna 25 is used for thetransmission and reception of transmission signals 35. Control lines 26arranged between the controller 21, the transmitter 2 and the receiver 8permit the selective operation of those devices to permit orderedbi-directional communication throughout the network 15. Each of thetransceiver modules 20 is connected to an appropriate device such as aLAN server 30, or devices 31, 32 and 33 respectively, which can bepersonal computers, printers, main frame computers, a connection to widearea network, and the like.

Referring to FIG. 3A, the simplest form of QAM is quadrature phase shiftkeying (QPSK) which provides a signal constellation 40 having fourconstellation points 41. In some instances, QPSK is known as 4 QAM.

FIG. 3B illustrates a 16 QAM signal constellation 42 which includessixteen constellation points 43 distributed in the manner shown.

FIG. 3C illustrates a 64 QAM signal constellation 44 which includessixty-four constellation points 45 also distributed in the manner shown.

FIG. 3D illustrates the effects of effects of AM/PM conversion on theideal constellation points 45 of one quadrant 46 of the constellation 44of FIG. 3C. When subjected to AM/PM conversion, an error vector 47 isapplied to each ideal constellation point 45 which causes thecorresponding real constellation point 48 to be displaced from its ideallocation 45. It will be appreciated from FIG. 3D, that the magnitude ofthe vectors 47 increase with the greater distance of the constellationpoints 45 from the origin 52 of the constellation 44. Importantly, indigital QAM systems, each constellation point represents a quantizedtransmitted value. Accordingly, schematically illustrated about each ofthe constellation points 45 is a home constellation box 50 such that anyconstellation point detected in that box will be interpreted as thecorresponding ideal constellation point 45. Accordingly, if the AM to PMvector 47 is small, the constellation point is not displaced too greatlyfrom the ideal location and no error results.

However, in this particular example, the furthest constellation pointfrom the origin 52 is subjected to a substantial error vector 47 suchthat the real constellation point 49 is displaced from its correspondinghome constellation box 50 and into a box 51 of an adjacentconstellation. Accordingly, such an error will be reproduced as an erroron demodulation of that data component. Further, it will be appreciatedthat AM/PM conversion results in the entire constellation rotating aboutthe origin 52 by the phase angle of the corresponding output power ofthe power amplifier. This is illustrated in FIG. 3E which shows how thephase angle changes as the output amplitude of the power amplifierincreases, to the extent that at high amplitudes, output compressionoccurs which causes large phase angle errors.

Further, even small errors (as opposed to the vector 47), can besignificant as they reduce the immunity the communication system has tonoise. If the transmitted symbols lie in their correct locations, thereis then maximum noise immunity. By shifting a symbol point half-waytoward any one (or more) boundary, the noise need only be half theamplitude otherwise required to cause a decision error.

One commonly used method for overcoming the problem of phase angleerrors, is to operate the RF power amplifier at approximately 10 dBbelow its rated output level. Accordingly, if a 1 watt transmitted powerlevel is desired, it is necessary to obtain and operate a 10 watttransmitter at a 1 watt level. In some instances, this can represent asubstantial cost difference over the required power output of acorresponding 1 watt transmitter. Further, the alternative, which is todevelop a highly linear 1 watt amplifier, can be just as expensive asthe cost of purchasing a 10 watt amplifier.

Another cause of phase errors is instability in the local oscillatorused in translating the received QAM signal from the RF carrierfrequency down to base band. Traditionally, this problem is overcomeusing an analog carrier recovery phase locked loop such as that shown ina QAM demodulator 60 of FIG. 4. There, an IF signal s(t) 61 isquadrature converted to base band using mixers 62 and 63. The downconverted signals are then passed through Arm filters 64 and 65respectively, which preferably perform an integrate and dump functionprior to passing the signal to analog-to-digital converters (ADC's) 66and 67, respectively. The ADC's 66 and 67 are used as the decisionmaking device, deciding which of the 2^(n) possible I and Q values asignal is, thus providing 2^(n) output bits per symbol on data outputlines 76a and 76b. Corresponding ideal signal values are thenregenerated by a digital-to-analog converters (DAC's) 68 and 69, so thatas far as the carrier loop is concerned, each ADC and DAC pair form aquantizer, shifting the signal amplitude to the nearest constellationpoint. A phase error signal is generated using two mixers 70 and 71, anda summer 72. The phase error signal is filtered in a loop filter 73 andthen used to drive a voltage controlled oscillator (VCO) 74 whichregenerates the carrier frequency. A quadrature phase shifter 75supplies the carrier to one of the mixers 63. The implementation of sucha demodulator for a symbol rate of 2 MHz is not readily achieved in lowcomponent count, low power technology. Further, the arrangement of FIG.4 does not provide any automatic gain control (AGC) function which isadvantageous to ensure optimal demodulation.

FIG. 5 illustrates a receiver configuration 80 intended for use with thepreferred embodiments and which incorporates an AGC function. An antenna81 is arranged to receive RF transmitted signals which are passedthrough a RF band pass filter 82 to reduce unwanted noise. The filteredsignal is then passed through a radio frequency amplifier 83 andsubjected to IF mixing in a mixer 84, also input from an IF oscillator85. The translated signal is then passed through an IF band pass filter86 prior being input to a voltage controlled amplifier (VCA) 87. Theoutput of the VCA 87 comprises an IF signal input s(t) 88 which is inputto spread-spectrum despreading box 79 which generates a symbolsynchronising pulse 98 and also passes the IF signal 88 to a QAMdemodulator 89 configured in accordance with an embodiment of thepresent invention. The demodulator 89 provides a data output 90 as wellas an AGC control voltage output 91 which supplies the VCA 87. Innon-spread spectrum applications, the despreading box 79 may besubstituted by a squaring device, and a band pass filter operating atthe symbol frequency.

The arrangement of FIG. 5 permits implementation of the demodulator 89using digital signal processing (DSP) which enables an equivalent of thecarrier recovery loop of FIG. 4 to be implemented at substantiallyreduced expense. A further advantage of such a configuration is that,through omitting the traditional VCO 74, the restriction on the VCOcapture range is removed, which aids acquisition and assists to preventfalse locking. Furthermore, the omission of the analog carrier recoveryloop of FIG. 4 permits the use of a fixed frequency local oscillator,preferably crystal based, that affords further cost reduction. Theability to use DSP also permits the implementation of a decision-aidedAGC system where the average value of the signal is not held constant.If the average power were used, the signal amplitude varies with thedata pattern being transmitted and so the response time of any such AGCsystem must be sufficiently slow in order to average over enough symbolsto attempt to obtain random data. Such a configuration is not suitablefor rapid acquisition in packet-type applications.

With a decision-aided AGC system, the gain error on a symbol-by-symbolcase is determined and can be used to control the VCA 87. Such anarrangement is prone to internal errors because, in order to work, thegain has to be nearly correct to start with. As such, there is no quickrecovery in decision-aided AGC systems.

A generalised configuration of the QAM demodulator 89 is shown in FIG. 6where the IF signal s(t) 88 is supplied to mixers 94 and 95 whichrespectively perform quadrature translation of the IF signal 88 tobaseband utilising a fixed frequency local oscillator 92 and aquadrature phase shifter 93. Each of the baseband signals is then passedthrough a corresponding integrator 96 or 97 which is reset using thesynchronising signal 98 which acts to dump the integrator outputs. Theintegrators 96 and 97 perform complementary functions to those of theArm filters 64 and 65 of FIG. 4. The filtered signals are then passedthrough the corresponding ADC's 99 and 100 to provide the in-phase (I)and quadrature (Q) signals that are respectively supplied to a digitalsignal processor (DSP) 101. The DSP 101 provides a demodulator dataoutput 90 together with an error signal 102 which is passed through aDAC 103 to provide the AGC control voltage 91.

The processing function required to be performed in the preferredembodiment by the DSP 101 is indicated in FIG. 7. There, the I and Qsignals are input to a complex multiplier 104 which effectively rotates(I+jQ) by exp(jΓ). The multiplier 104 has two outputs 105 and 106 whichare each provided to respective quantizers 107 and 108 whichrespectively provide the data output components 90_(I) and 90_(q)representing the constellation points.

Having thus performed QAM demodulation, the remainder of the function ofFIG. 7 is devoted to phase correction and gain control. Firstly, withreference to phase correction, each of the quantizers 107 and 108 has acorresponding further output 119 and 120 which is supplied to acorresponding mixer 109 or 110 also connected to the output 106 or 105respectively. The mixers 109 and 110 output on an adder 111 whichprovides a phase error value 112. The phase error value 112, whichoccurs for each symbol of the constellation, is then processed to obtainthe actual phase rotation angle of the constellation, as previouslyillustrated in FIG. 3D. This is performed by supplying the phase errorvalue 112 to an accumulator 115 which outputs to a constant multiplier116, the output of which is then added in a summer 114 with a valueobtained from a constant multiplier 113, also input with the phase errorvalue 112. The output of the summer 114 represents the phase error inthe frequency domain, which is then applied to an accumulator 117 whichconverts the value into the time domain to provide the actual phaseerror angle θ 118 which is returned to the complex multiplier 104 toform a feedback loop. In this manner, a digital equivalent of thecarrier recovery loop of FIG. 4 is achieved without the use of a voltagecontrolled oscillator and with a configuration that can be implementedsolely with digital processing. Importantly, the digital value of phaseerror is used to form the carrier recovery loop which effectivelyperforms a complex multiplication.

With reference to gain error correction, each of the quantiser outputs119 and 120 are supplied in turn to corresponding mixer and squaringcircuits 121, 124, and 122, 125 respectively. The output of each of themixers 121 and 122, which corresponds to the amplitude of the error inthe quantising process, is supplied to a summer 123. The outputs of thesquaring circuits 124 and 125 are also supplied to a summer 126. Thesummers 123 and 126 each supply a differential summer 127 which providesa gain error value 128. The gain error value 128 is then supplied to anaccumulator 129 which provides the actual gain error 102. The error 102is then passed through the DAC 103 to provide an analog AGC controlvoltage 91.

The demodulation function 101 depicted in FIG. 7, whilst permittingdesign flexibility and accurate correction for gain and phase errors, issufficiently complex in processing so as to prevent a desired symbolrate of 2 MHz to be achieved using lowcost DSP units.

However, as depicted, the DSP function 101 is arranged into an upperblock 130 and lower block 131. Importantly, the upper block 130 providesa memory-less structure having (Ni+Nq+Nt) inputs and (2Nd+Ne+Na)outputs. In view of the demodulation function being performed upondigital values, and due to a clearly identifiable relationship betweenthe number of inputs and outputs, a direct correlation between input andoutput values can be obtained suitable to permit implementation of thefunction 101 as a look-up table that can be realised using read-onlymemory (ROM) devices, when suitably programmed. In such animplementation, the important parameter of note is the actual size ofthe ROM, and this depends on the number of bits of resolution requiredfor the various signals, which can be calculated empirically. However,importantly, for a 64 QAM system, good performance can be obtained using1 megabyte ROM. Further, optimisation of the DSP function can reducethis requirement to 256 kilobytes. Also, at a proposed symbol rate of 2mega symbols per second, the ROM access time is of the order of 500nanoseconds, which is easily met with current technology.

The lower block 131 contains memory devices (the accumulators 115, 117and 129) and as such is not suitable for implementation as a look-uptable. The lower block 131 however can be implemented in a DSP device,or alternatively as part of a VLSI integrated circuit. Further, thememory requirements of the accumulators 115, 117 and 129 are not largeand such devices only have to operate with cycle times of the order of500 nanoseconds. Further, the gain constants "a" and "b" of themultipliers 113,116 can also be arranged in powers of two, therebyfacilitating implementation using shift registers.

FIG. 8 illustrates a QAM demodulator 134 configured to implement thedemodulation function of the arrangement of FIG. 7. A ROM 135 isprovided which accepts digital I and Q signals each having N bits ofdata. A data output 90 provides 2N bits of demodulated data comprising Iand Q components, 90_(I) and 90_(Q). The ROM 135 also outputs the gainerror value 128 and the phase error value 112 to a VLSI device 136 whichincorporates components corresponding to those of the lower block 131 ofFIG. 7. Those components output the phase angle error θ 118 which isreturned as an input in a feedback loop to the ROM 135, which performsthe carrier recovery operation. The provision of the gain error value128 effectively tracks the phase offset for each amplitude of thereceived constellation and thus can be used to improve the Bit errorrate (BER). of the demodulation system. In this embodiment, the AGCvalue is determined on a symbol-by-symbol basis from the gain errorvalue 128 produced from the ROM 128. Essentially, this configurationprovides for a determination of the symbol, and from that, adetermination of the gain error, and thus differs from the "average" and"decision-aided" approaches referred to above. The arrangement thus actsto compensate for any AM/PM distortion in the power amplifier 4 of thetransmitter 2.

It is to be noted that FIG. 8 does not illustrate any timing signalsneeded to control the flow of data around the ROM 35, such as latchingof the ADC data, when to operate the accumulators 115,117,129, and whenthe output data 90 is valid. However, such details would be wellunderstood by those skilled in the art on the basis of current existingtechnology. Such extra logic can be, and is preferably developed, aspart of the VLSI device 136. The ROM 135 and VLSI device 136 updatetheir states once per symbol, and for a 2 mega symbol per second system,that is every 500 nanoseconds. Thus, following the ADC devices 99 and100 of FIG. 8, a minimum of only two integrated circuit devices would berequired to complete such a 64 QAM demodulator.

An additional advantage of this technique is that other signalconstellations can be handled apart from the classic square (8×8) 64 QAMconstellation, illustrated in FIG. 3C. No other circuit modificationsare required, only re-programming of the ROM 135. Accordingly, if aconstellation has better properties that requires extremely complicatedprocessing to determine the transmitted symbol, the gain and phaseerrors for the AGC and carrier recovery loops respectively can bedetermined offline and the results placed in a ROM.

The demodulation of a QAM signal requires tracking the phase of thereceived signal. Frequency offsets between the transmitter and receivercause a regular phase change each symbol. The phase changes caused bythese frequency offsets can be measured and tracked by the carriertracking loop if the phase change per symbol is less than that needed torotate a symbol into the decision area of a neighbour in theconstellation. For QPSK, this is 45 degrees per symbol (whichcorresponds to a frequency offset of 250 kHz for a symbol rate of 2MHz). For 64 QAM, the maximum change that can be tolerated is of theorder of 8 degrees (the exact number depending on the data pattern)which corresponds to a frequency offset of 44 kHz for a symbol rate of 2MHz. If the RF carrier frequency is 2.45 GHz then the maximum frequencyoffsets that could be tolerated would be about 102 ppm for QPSK, 18 ppmfor 64 QAM, and similarly about 42 ppm for 16 QAM.

Low cost, free running, crystal oscillators typically give ±50 ppmaccuracy, allowing for manufacturing tolerances and temperature andaging effects. Accordingly, with such devices, QPSK using the preferredembodiments can be supported. However, with such low cost devices, 16QAM and 64 QAM would not appear directly supportable.

Notwithstanding the above, carrier recovery can be performed using QPSK,which can then permit switching to either 16 QAM or 64 QAM. Such ascheme is best used where the data is transmitted in packets.

As previously discussed, practical RF power amplifiers have AM/PMconversion, and also AM-AM conversion. The latter is simply gaincompression and can be corrected at the transmitter using a variety oftechniques, for example predistortion, being one of the simplest toimplement. Further, as indicated in FIG. 2, as the transmitter isactually part of a transceiver, the corresponding receiver can be usedto monitor the transmitter output and to correct the transmittedamplitudes.

The manner in which the ROM-based demodulator 134 of FIG. 8 compensatesfor AM to PM conversion is best understood with reference to FIG. 3D.The ROM 135 takes the received signal (I+jQ) and rotates it by the angleθ before making a decision on the constellation point received. Themechanism by which the AM to PM conversion is removed is to add a phaseoffset to θ prior to it being used to rotate the receive signal. Thisoffset is a function of the signal amplitude.

The method of compensating for the AM to PM is to:

(a) determine the signal level;

(b) determine the phase offset using a look-up table; and

(c) add the phase offset to the current estimate of signal phase.

The accuracy of the compensation required depends on the accuracy withwhich the received constellation must be constructed. This is a functionof both the level of the modulation and the required error rate.

The proposed method of measuring the signal amplitude is to incorporateit into the lookup table. If the AM to PM results in a distortion whichcan be estimated once, to an adequate accuracy for all transmitterslikely to be received, then this distorted constellation can be directlyprogrammed into the ROM and no further compensation is necessary.

Otherwise the amplitude signal must be used to select a phase offsetprior to the decision process. Such an arrangement is shown in FIG. 9,where a QAM processor 140 is shown to includes an amplitude decisioncircuit 141 which receives the value of (I+jQ) and which controls aswitch 144 to select one of four basic phase offsets 142. The selectedoffset is added in a summer 143 to the value output from the VLSI device136 to give the desired phase error value θ, which is used by the ROM inthe previous manner.

Thus, a demodulator including the QAM processor 140 can compensate theAM/PM distortion if it is a known quantity.

However, in practice, this must be estimated from the received signal.Preferably, this is done is by transmitting a fixed header signal on thedata packet, and the AM/PM distortion is determined using the fixedheader signal. Initial phase acquisition is preferably performed usingQPSK, which essentially corresponds to the extreme corner constellationpoints in higher order systems.

Simulations indicate that using 64 QAM, if a channel raw Bit Error Rate(BER) of 10⁻⁵ is required, then the Eb/No needed by an ideal demodulatoris 17.7 dB. Using the ROM demodulator proposed it is possible to achievethis error rate at Eb/No of 19.4 dB (a degradation of 1.7 dB, which canbe reduced by increasing the size of the ROM). Simulating the effect ofa practical power amplifier indicates that operating at 6 dB backoffgives a further degradation of 1 dB. (Hence a 4 watt amplifier is neededto transmit 1 watt).

Typical simulator results using the compensated demodulation arrangementof FIG. 9 are that if the transmitter is operated at 1 dB backoff, andthe amplitude compression is corrected at the transmitter, then a phasecompensation system compensating the four largest constellation pointscauses negligible extra degradation (ie. at 1 dB backoff usingcompensation the total degradation is 1.7 dB). Thus using compensation,to transmit 1 watt, only a 1.25 watt amplifier is required. As a result,it may be possible to operate with less backoff.

Wireless LAN environments have communications links that have greatlyvarying path loss characteristics, both due to fixed factors (eg.location relative to the base transmitter) and variable factors (eg.movement of office staff). It is possible to operate over a path withgreater loss by decreasing the transmitted data rate. In the simpleembodiments, the data rate can be set low enough so that communicationis possible over paths with the maximum likely loss. This does notresult in optimum use of the available spectrum. It is important in suchan environment, that the spectral resources be optimally located tomaximise network efficiency. This is achieved by:

using a low data rate for addressing and paging information, whichmaximises the reliability of this channel for minimal allocation ofresources; and

using the highest data rate possible on a channel, which often requiresdata rate adaption.

The QAM ROM-based demodulators of the described embodiments can beconfigured to switch instantly from one modulation system to another, bysimply increasing the size of the ROM to include constellationinformation for other systems. For a 64 QAM demodulator it thereforepossible, by doubling the size of the ROM, to demodulate 64 QAM, 16 QAMand QPSK. This would give relative data rates of 3:2:1 and relativeperformances in noise of 0:+5.5 ds:+11.9dB (allowing for Eb/Noimprovement and the peak power limitation of the system). That is, bydropping the data rate by a factor of 1/3, the communications link cannow operate with 11.9 dB more path loss at the same error rate.

Further, the embodiments can also be arranged that the values output forthe phase (112) and amplitude (128) errors are appropriately scaled toenable the same carrier recovery circuit to be used in the VLSI. Noadditional carrier acquisition is needed following the switch 144.

The preferred embodiment when implemented for wireless LAN has a datastructure as shown in FIG. 10. The data is transmitted in packets 210,having a QPSK preamble 202 for carrier acquisition, and a QPSK header204 containing addressing, paging and other control channel information.A short coded sequence 206 is provided for AM/PM adaption (which ispreferably QPSK, and so could also carry information). An optional datarate switch 208 can then be provided before the information data 210 ofthe packet 200. The existence of the data rate switch 208 is required tobe previously flagged, preferably in the header 204, and can be used toswitch from QPSK (as preferred for the preamble 202, header 204, andlearning sequence 206) to either 16 QAM or 64 QAM used in the data 210.Such a data rate switch is readily implemented in the preferredembodiments because, for each coding scheme (QPSK, 16QAM, 64QAM), thegain and phase values output from the ROM all fall within the sameboundaries and thus hardware adaption is not required.

In order to optimize the described embodiments, it is desirable toreduce the size of the ROM 135. It is generally difficult to reduce thenumber of I and Q bits used in a QAM demodulation however, the number ofθ bits can be reduced by trading them with some preprocessing. Byexamining the most significant θ bits the following is noted:

MSB--the most significant θ bit corresponds to a rotation of 180°;

MSB-1--the next most significant bit corresponds to a rotation of 90°;

MSB-2--the next most significant bit corresponds to a rotation of 45°; .. . and so on.

These bits can be removed from the look-up table addressing if theirfunctionality can be provided elsewhere in the demodulation system. TheMSB can be removed and replaced either by preprocessing, for example byinverting the I and Q data when the MSB is set, or by post-processingthe data, for example by replacing the demodulated symbols with those180° away. A similar operation can remove the next most significant bit.The removal of higher bits is complex due to the lack of symmetry of theconstellation under rotations less than 90°. Implementation of therotation for these bits is likely to be performed simplest using thelookup table. Two arrangements which reduce the ROM requirements from 1megabyte to 256 kilobytes for a 64 QAM demodulator are shown in FIGS. 11and 12.

In FIG. 11, a demodulator 160 is shown in which components previouslyseen and similarly numbered perform corresponding functions. A QAMprocessor 161 is provided which incorporates a VLSI device, a first partof which (Part A) 163 is provided to receive the I and Q channels fromthe ADC's 99 and 100. Part A VLSI 163 forms rotations by 0°, 180° and±90°. This requires sign changes and exchanging I and Q data. The datais then output to a reduced size ROM 162 which provides a data output 90in the previous manner. The ROM 162 also supplies gain and phase errorsignals 112 and 128 to a Part B VLSI 164 which implements thecorresponding function seen in FIG. 8. The only variation is that the θvalue, comprising 7 bits, is divided between the VLSI Part A 163 (2bits) and the ROM 162 (5 bits). This preprocessing technique requiresminimal hardware as the rotations required provide simple sign changesand data interchanges. The processing can be made more simple if the ADCdata is presented in a signed binary form.

In the configuration of FIG. 12, a demodulator 170 is provided whichincorporates a QAM processor 171. The processor 171 incorporates areduced sized ROM 173 which outputs to a VLSI device 174 which receivesgain and phase error signals 112 and 128 in the previous manner andprovides a θ and AGC values in the manner previously described. The QAMprocessor 171 performs post-processing using a second ROM 175 interposedbetween the data output 90 and the ROM 173 and which also receives 2bits of the 7 bit θ signal, the remaining 5 bits being input to the ROM173. Post-processing performs rotations by 0°, 180° and ±90°. The mainROM 173 is programmed as before except that the 5 bits of rotation usingthe angle θ only range from 0 to 90°. Decisions made from the ROM 173need rotation to correct quadrant. This is performed by the secondsmaller ROM 175 which only requires 2 bits of θ input, thereby beingable to identify the four quadrants of the constellation.

The arrangements described above permit configuration of a wireless LANoperating at a carrier frequency of about 2.45 GHz, and capable oftransmitting data at a rate of 10 Mbps using a 64 QAM system.

The foregoing describes only a number of embodiments of the presentinvention and modifications, obvious to those skilled in the art can bemade thereto without departing from the scope of the present invention.

What is claimed is:
 1. A demodulation arrangement comprising:a mixingcircuit for deriving in-phase (I) and quadrature (Q) signals in adigital form from a received signal; a look-up table for outputting adata output value and a corresponding phase output value, bothselectable using said I and Q signals; and a phase processor arranged toreceive said phase output value and to calculate a phase angle value ofsaid received signal in a time domain by accumulating a phase anglevalue in a frequency domain; wherein said phase angle value in the timedomain is further input to said look-up table to supplement said I and Qsignals for selection of at least said data output value.
 2. Anarrangement as claimed in claim 1, wherein said I and Q signals providea determination of a level of said received signal, said phase outputvalue being derived from said table by association with said level. 3.An arrangement as claimed in claim 1, wherein said phase processor addssaid phase output value to a current estimate of signal phase to providesaid phase angle value.
 4. An arrangement as claimed in claim 3, whereinsaid current estimate of signal phase is determined by a firstaccumulator whose weighted output is summed with a weighted version ofsaid phase output value, the summation being subsequently summed in asecond accumulator over a plurality of symbols to provide said phaseangle value.
 5. An arrangement as claimed in claim 4, further comprisingan angle selector configured to provide a basic phase angle amount towhich said phase angle value is added prior to being input to saidlook-up table.
 6. An arrangement as claimed in claim 5, wherein saidangle selector comprises an amplitude decider configured to receive saidI and Q signals and therefrom determine a magnitude of said receivedsignal, said magnitude selecting one of a plurality of said basic phaseangle amounts corresponding to said magnitude.
 7. An arrangement asclaimed in claim 1, wherein said look-up table further comprises, and isconfigured to output, a gain output value selectable using said I and Qsignals, said gain output value representing a gain error on asymbol-by-symbol basis.
 8. An arrangement as claimed in claim 7, furthercomprising a third accumulator for receiving said gain output value andaccumulating same over a plurality of symbols, an output of whichrepresenting a gain error of said received signal.
 9. An arrangement asclaimed in claim 8, further comprising a gain controllable amplifierconfigured to provide said received signal to said mixing means, saidamplifier having a gain control input with said gain error.
 10. Anarrangement as claimed in claim 1, wherein said look-up table comprisesa read-only memory (ROM).
 11. An arrangement as claimed in claim 10,wherein said ROM is implemented in a first single package electronicdevice, and said phase processor is implemented in a second singlepackage electronic device.
 12. An arrangement as claimed in claim 11,further comprising a third accumulator configured within said seconddevice for receiving said gain output value and accumulating same over aplurality of symbols, an output of which representing a gain error ofsaid received signal, andwherein said look-up table further comprisesand is configured to output, a gain output value selectable using said Iand O signals, said gain output value representing a gain error on asymbol-by-symbol basis.
 13. An arrangement as claimed in claim 11,wherein said phase processor further comprises a pre-processorconfigured to receive said I and Q signals and a part of said phaseangle value, a remainder of said phase angle value being input to saidlook-up table together with associated I and Q outputs of saidpre-processor, said pre-processor being configured compensate for bulkphase angle values whereby said look-up table need then only compensatefor complementary discrete phase angle values.
 14. An arrangement asclaimed in claim 11, wherein said phase processor further comprises apost-processor configured to receive said data output values from saidlook-up table and a part of said phase angle value, a remainder of saidphase angle value being input to said look-up table, wherein saidpost-processor provides associated outputs of said demodulator and isconfigured to compensate for bulk phase angle values whereby saidlook-up table need then only compensate for complementary discrete phaseangle values.
 15. An arrangement as claimed in claim 13, wherein saidlook-up table which need only compensate for the complementary discretephase angle values is of reduced capacity compared with said read-onlymemory (ROM) look-up table.
 16. An arrangement as claimed in claim 14,wherein said post-processor comprises a read-only memory.
 17. A radiofrequency receiver comprising an arrangement as claimed in claim
 1. 18.A radio frequency transceiver comprising a receiver as claimed in claim17.
 19. A communications system comprising a plurality of radiofrequency transceivers as claimed in claim
 18. 20. An apparatuscomprising:a read-only memory (ROM) in which data values and phasevalues are stored for selection by I and Q input data; a phase processorfor determining, from a plurality of said phase values, a current phaseangle of said I and Q input data in a time domain by accumulating aphase angle in a frequency domain; and a feedback path by which saidcurrent phase angle in the time domain is input directly to said ROM formodifying said I and Q values prior to said selection.
 21. Ademodulation apparatus comprising a read-only memory (ROM) configuredfor decoding symbol constellation points from a received signal, whereinsaid ROM outputs a phase output value, and a feedback path is formedfrom the phase output value and returning to said ROM, said feedbackpath implementing a digital carrier recovery loop incorporating acomplex multiplication function for determining a Phase angle value ofthe received signal in a time domain by accumulating a phase angle in afrequency domain.
 22. A method of demodulating a modulated signal, saidmethod comprising the steps of:(a) deriving in-phase (I) and quadrature(Q) signals from said signal; (b) obtaining a decoded data output valueand a phase output value from a (first) look-up table corresponding tosaid I and Q signals; (c) determining a phase angle value of said signalin a time domain from said phase output value by accumulating a phaseangle in a frequency domain; and (d) using said phase angle value inconcert with said I and Q signals in the obtaining of at least saiddecoded data output value.
 23. A method as claimed in claim 22 whereinstep (b) comprises using said I and Q signals as addresses in said(first) look-up table of data output values and phase output values forselection of said decoded data output value and said phase output value,and step (d) comprises using said phase angle value as a further addressto said first look-up table.
 24. A method as claimed in claim 23 whereinstep (c) comprises performing a predetermined processing of said phaseoutput value external to said first look-up table.
 25. A method asclaimed in claim 24 wherein step (c) further comprises the steps of:(ca)detecting an amplitude of said I and Q signals; (cb) selecting one of aplurality of phase angle offsets using said detected amplitude; and (cc)combining the selected phase angle offset with a processed phase anglederived from said phase output value to obtain said phase angle value.26. A method as claimed in claim 25 wherein steps (ca) and (cb)comprises inputting said I and Q signals into a (second) look-up tableto select said phase angle offset from a plurality of phase angleoffsets retained in said second look-up table, and step (d) comprisesinputting said phase angle offset and said processed phase angle incombination to said first look-up table.
 27. A method as claimed inclaim 24 wherein step (b) comprises the further steps of:(ba) inputtingsaid decoded data output value and at least a portion of said processedphase angle into a (third) look-up table to select a specific dataoutput value from said third look-up table.